Delay-Locked Loop with Dynamically Biased Charge Pump

ABSTRACT

A delay-locked loop, including a phase detector configured to receive two signals, one of the signals being delayed relative to the other of the signals, the phase detector having an UP output and a DOWN output. The delay-locked loop also includes a charge pump system operatively coupled with the phase detector, the charge pump system including (1) a charge pump configured to be responsive to assertion of actuating signals from the UP output and the DOWN output so as to control pumping of charge from the charge pump system, such pumped charge being usable to control a delay line carrying one of the two signals, so as to control relative delay occurring between the two signals; and (2) a feedback control loop configured to dynamically adjust at least one bias signal at the charge pump so as to minimize net charge pumped from the charge pump system during simultaneous assertion of actuating signals from the UP output and the DOWN output.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation of U.S. patent application Ser. No.13/421,274, filed Mar. 15, 2012 and entitled DELAY-LOCKED LOOP WITHDYNAMICALLY BIASED CHARGE PUMP; which is a continuation of U.S. patentapplication Ser. No. 12/634,596, filed Dec. 9, 2009 and entitledDELAY-LOCKED LOOP WITH DYNAMICALLY BIASED CHARGE PUMP; which is acontinuation of U.S. patent application Ser. No. 11/347,835, filed Feb.3, 2006, now U.S. Pat. No. 7,634,039, and entitled DELAY-LOCKED LOOPWITH DYNAMICALLY BIASED CHARGE PUMP; which in turn claims the benefitunder 35 U.S.C. §119(e) of the U.S. Provisional Patent Application Ser.No. 60/649,948, filed Feb. 4, 2005 and entitled MASTER/SLAVE ANALOG DLLWITH LOW JITTER, the entire disclosures of which are incorporated hereinby reference in their entirety for all purposes.

TECHNICAL FIELD

This disclosure relates to Delay-Locked Loops (DLLs), and in particularto the reduction of jitter, duty cycle distortion, and static phaseoffset in DLLs.

BACKGROUND

DLLs have many applications. For example, frequency-multiplying DLLs canbe used to generate high frequency clocks from low frequency clocks.Deskew DLLs can be used to phase align a distributed clock to areference clock. Master/slave DLLs can be used to delay arbitrarysignals by precise fractions of a clock period.

Because DLLs often must produce large delays, they may also addproportionally large jitter (random variations in delay) to the signaldelayed. Such jitter is usually undesirable, and so techniques forreducing jitter are valuable.

Another result of the large delays in the delay line is significantdifferences in the propagation time of low-to-high versus high-to-lowedges. These differences show up as duty cycle distortion, or moreaccurately named pulse width distortion, at the output of the DLL, andare also undesirable.

In addition, DLLs often suffer from inaccuracies in the control systemthat aligns the delayed reference clock edge with an undelayed edge. Anyinaccuracy in this alignment is called static phase offset. Static phaseoffset leads to inaccuracies in the delayed edge timing, which areundesirable.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram of a delay-locked loop (DLL) system accordingto the present description.

FIG. 2 is a block diagram of a DLL application involving slave delaylines.

FIG. 3 is a schematic diagram of a delay line.

FIG. 4 is a schematic block diagram showing a charge pump systemaccording to the present description.

FIG. 5 is a schematic circuit diagram of a charge pump embodiment thatmay be employed with the charge pump system of FIG. 4.

FIG. 6 is a schematic circuit diagram of a charge pump replicaembodiment that may be employed with the charge pump system of FIG. 4.

FIG. 7 is a schematic circuit diagram of an amplifier embodiment thatmay be employed with the charge pump system of FIG. 4.

FIG. 8 is a schematic depiction showing one delay element embodiment ofa delay line that potentially includes multiple delay elements.

FIG. 9 is a schematic depiction of two delay elements within a delayline, depicting the connections between the delay elements and aflip-stack implementation of a NAND gate.

FIG. 10 is a schematic depiction of a further embodiment of a delay lineaccording to the present description.

FIG. 11 is a schematic depiction of a current dump, a circuit fordetecting an edge propagating within a delay line and conducting currentdepending on the presence or absence of such an edge.

FIG. 12 depicts another embodiment of a DLL system according to thepresent description, including a secondary control loop for providingdelay adjustments for an opposite edge type.

FIG. 13 depicts an alternate implementation of a portion of the DLLsystem shown in FIG. 12.

DETAILED DESCRIPTION

FIG. 1 depicts an embodiment 20 of a delay-locked loop (DLL) accordingto the present description. DLL 20 includes delay lines 22 and 24,configured to receive an input reference signal, such as REFCLK. Outputs26 and 28 of the delay lines 22 and 24 are respectively coupled to phasedetector 30. Phase detector 30 may include UP and DOWN outputs 32 and34, as will be explained in more detail, which may be applied as inputsto control operation of charge pump system 40. Charge pump system 40pumps charge (e.g., provides a momentary output current) based onapplication of the UP and/or DOWN signals. The pumped charge is used toprovide feedback to control delay lines 22 and 24, optionally withadditional processing by filter 42 and bias generator 44, in order tocontrol operation of the delay lines. When operating in lock, thecontrol system of the DLL described keeps the delay of individual delayelements fixed through variations in the operating environment, such astemperature and voltage fluctuations. According to an example discussedbelow, the fed back input to the delay lines is a voltage or currentwhich controls how much the respective delay lines delay the incomingreference signal. Typically this voltage or current is implemented as abiasing signal applied to delay elements in the delay line, so as tocontrol the speed of the delay elements.

FIG. 2 shows an exemplary embodiment in which one or more slave delaylines 60 and 62 are provided, and controlled based on output from a DLL,such as the DLL of FIG. 1. In such a scheme, the DLL of FIG. 1 may beconsidered to include master delay lines (e.g., lines 22 and 24), whilethe specifically depicted delay lines in FIG. 2 would be consideredslave delay lines. Any number of slave delay lines may be employed toprovide varying delay controlled by the closed-loop controlled delay ofthe master loop. In particular, the slave section includes one or moredelay lines which use controls from the master section to delay anarbitrary signal (e.g., the signals shown applied through the slavedelay lines 60 and 62) by some precise fraction of the reference clockperiod. In different implementations, the fraction may be fixed or maybe programmable. For example, in Double Data Rate (DDR) memoryapplications, it is often desired that the slave delay lines have adelay that is one quarter the master delay which is locked to areference clock period.

DLL delay lines, whether master or delay, typically are composed ofmultiple delay elements. The amount of signal delay provided in a givenline may be controlled by varying the delay of the individual delayelement(s) and/or by selecting a number of delay elements which will beoperative or activated. In other words, the more delay elements that areactivated, the longer the signal will be delayed, because it travelsthrough a greater number of delay elements before being output.Additionally, or alternatively, the speed or delay of various types ofdelay elements may be controlled by varying control voltages or currentsapplied to the delay elements (e.g., such as a regulated supply voltagethat drives logic gates in the delay elements).

Varying the number of delay elements can be used to lock the delay linebased on the reference clock and/or to replicate a scaled version of adelay. FIG. 3 shows an example of a delay line 80 including two delayelements 82 and 84. Two elements are included for clarity of explanationonly; it should be understood that delay lines commonly include manymore elements, depending on the particular application or intended use.Continuing with the present example, in each delay line element, thelower multiplexor leg may be called the “feed-forward” leg, and theupper multiplexor leg may be called the “wrap” leg.

Decoder 86 may be arranged so that, from the left, the multiplexors ineach delay element are switched to their feed-forward legs until onemultiplexor is switched to its wrap leg. As explained below, decoder 86provides control over how many delay stages (i.e., elements) areactivated within delay line 80, and thus over how much delay isintroduced by the delay line.

Continuing with the example of FIG. 3, the signal to be delayed travelsfrom input 88, through potentially many buffer stages to the right,until it comes to the multiplexor which passes it through the wrap leg.The buffer stages passed up to this point may be referred to as the“input chain”. At that point it passes through potentially many bufferstages until it passes to the output. The buffer stages after themultiplexor which wraps the edge are the “output chain.” The controlsignal from decoder 86 thus controls how many buffers the signal to bedelayed must travel through to reach output 90.

The master delay lines and other components of FIG. 1 typically areconfigured so that the difference in delay between delay lines 22 and 24is one clock period of the reference signal. This includes controllingthe relative delay introduced by delay lines 22 and 24. As indicatedabove, this delay adjustment can be done through adjusting the delay ofeach element, and/or by varying the number stages in the delay line.

Continuing with the figure, delay line 22 is adapted so that it delaysthe reference clock signal by one cycle, as indicated in the figure(360°). This may be achieved by beginning with an initial approximationand subsequent adjustment, by empirically selecting a delay amount atdesign time, by selectively trying different numbers of delay elements,etc. Delay line 24 is adapted so that it delays the signal by someminimum amount. In practice, delay line 24 provides some minimum(non-zero) amount of delay, such that this amount typically is accountedfor in the delay of delay line 22. In other words, whatever the minimumdelay of line 24, delay line 22 typically is adapted to provide anadditional delay in the amount of one clock period.

Although the present examples primarily discuss a master loop employingtwo delay lines, other configurations may be employed. For example,instead of a 0° delay line (line 24), the reference signal (REFCLK) maybe applied directly to the phase detector. In another example, a matchcell is employed in addition to or instead of the 0° line. Indeed,except where otherwise indicated, the present disclosure is applicableto a variety of settings in which two signals are applied to a phasedetector for purposes of generating feedback control to control delayoccurring between the two signals.

The advantage of using two delay lines in the master loop is that thereference clock period is locked to the delay difference between them.This makes it possible to program the slave delay line to delay a signalby a precise fraction of the reference cycle, relative to a match cell.

Phase detector 30 typically is configured to compare each reference edge(from delay line 24) to the previous reference edge delayed by one cycle(from delay line 22). If the delayed reference edge is early, the delayline is too fast, so the phase detector produces a proportionally longDOWN pulse. If the delayed reference edge is late, the delay line is tooslow, so the phase detector produces a proportionally long UP pulse. TheUP and DOWN outputs 32 and 34 cause the amount and sign of the chargepumped to vary, which in turn causes the delay elements in lines 22 and24 to run faster or slower, so as to bring the current and delayed edgescoming out of the delay lines into alignment. For example, the UP signal32 may be pulsed when the delayed edge is lagging, thereby increasingthe speed of the delay elements and reducing the relative delay of thedelayed edge. Conversely, the DOWN signal 34 may be pulsed to increasedelay in the event that the delayed edge is ahead of the current edge.

Typically, phase detector 30 is configured to pulse both UP and DOWNoutputs when current and delayed edge are aligned, or nearly aligned(e.g., in lock). For example, a range of offsets around absolute lockmay be selected in which both UP and DOWN outputs are pulsed. Asdiscussed below, this may eliminate or reduce a “dead band” in which thesystem does not tightly control or maintain lock when the current anddelayed edges are in lock or nearly in lock. When a dead band ispresent, there may be jitter or slightly varying phase offsets betweenthe edges that are to be aligned. In many cases, this diminished controlis undesirable.

Charge pump system 40 receives the UP and/or DOWN outputs from phasedetector 30 and produces a pumped charge output proportional to thepulse width on the UP or DOWN outputs for loop filter 42. Otherimplementations may include producing a fixed charge rather than aproportional charge in response to phase differences between thereference and delayed signals output from delay lines 22 and 24.

Loop filter 42 may be implemented with a loop filter capacitor thatstores the current delay value as the control voltage. This voltage isproportional to the time integral of the error charge signal from thecharge pump.

Bias generator 44 may be adapted to create a regulated supply voltageVVdd for driving the delay lines 22 and 24, and other system components.Typically the regulated supply voltage is less than the block supplyvoltage, and is constant and independent of changes to the block supplyvoltage. This regulated supply voltage is used to power the master delaylines (e.g., lines 22 and 24). According to one example, bias generator44 receives the control voltage and creates a regulated supply voltage,related or equal to the control voltage, that is used to directlycontrol the delay of the master delay lines. According to anotherexample, a constant supply voltage is based on some other constantvoltage, such as a band gap voltage, and a separate method is used tocontrol the delay of the delay line based on the control voltage.

According to another embodiment, a digital DLL scheme may be employed,in which a digital up/down counter is employed instead of analog chargepump system 40 and loop filter 42. The counter is responsive to pulsesfrom phase detector 30, so as to digitally control the bias generatorand/or vary the number of stages in the delay line (with or withoutusing a bias generator).

Assuming that the delay line has sufficient delay range, and that thecontrol loop is properly damped, the master loop described in connectionwith FIG. 1 is adapted to converge on a state in which delay line 22delays the reference clock by one reference cycle, or in the case ofnon-negligible delay in delay line 24, by one reference cycle more thanthe delay occurring in line 24.

It will be appreciated that the described DLL system may be employed ina variety of settings. Some examples, as above, involve a delay betweenlines 22 and 24 of one reference cycle. Other exemplary settings may beconfigured for different amounts of delay.

The following are exemplary applications for the DLL of FIG. 1:

Example 1: In a deskew DLL, delay line 22 delays by less than one cycle,with its output feeding a clock distribution. One end point of thatclock distribution may be the input to phase detector 30.

Example 2: In a frequency-multiplying DLL, the output of delay line 22is multiplexed back into its own input. In such an exemplary topology, aclock pulse may be circulated through the delay line multiple timesbefore arriving at phase detector 30.

Example 3: In a master/slave DLL (referring also to FIG. 2), the masterDLL of FIG. 1 may be used to drive one or more slave delay lines. In theanalog example of FIG. 1, a control voltage, either from loop filter 42or bias generator 44, is applied from the master DLL to drive one ormore slave delay lines 60, 62, etc. In this example, the control voltagecontrols the speed of the delay elements in the slave delay lines (thuscontrolling the delay). Additionally or alternatively, a control signalmay be employed to vary the number of operative delay elements in theslave lines. In either case, a digital control of the slave delay linecan then select a delay equal to a precise fraction of the master loop'sdelay.

Referring now to FIGS. 4-7, an exemplary embodiment of charge pumpsystem 40 will be described. Charge pump system 40 may include adifferential amplifier 102 and a charge pump replica stage 104, as wellas the actual charge pump stage 106. Charge pump 106 includes UP andDOWN charge pump inputs (e.g., received from phase detector 30), and acharge pump output, which may be stored as a control voltage Vcnt1 via acapacitor, loop filter or other mechanism. Replica charge pump 104includes a replica charge pump output. In FIGS. 5-7, exemplary relativedevice widths are indicated adjacent the gate of the depictedtransistors. It should be appreciated that various alternate devicesizes and configurations are possible, and that the present depictionsare exemplary only. The depicted sizes/configurations may providecertain advantages. For example, the depicted sizes/configurations mayensure that the NMOS differential pair devices in the amplifier stay insaturation for all possible Vctr1 voltages. These ratios are highlightedin the charge pump figures and will be explained below.

Both replica charge pump 104 and charge pump 106 include first biasinginputs tied to the same value, designated in the figures as Vbp. Vbp mayalso be referred to as the PMOS bias voltage of the exemplary chargepump system 40. In typical exemplary operational settings, Vbp isslightly below (e.g., 200 mV) a supply or block voltage Vdd of thesystem, or other upper rail voltage or reference.

In addition, replica charge pump 104 and charge pump 106 include secondbiasing inputs, which are also tied to the same value, designated in thefigures as Vbn. Vbn may also be referred to as the NMOS bias voltage ofthe exemplary charge pump system 40. In typical exemplary operationalsettings, Vbn is slightly above (e.g., 200 mV) a substrate voltage Vss,or other ground or lower rail or reference voltage.

In the depicted example, the charge pump output (e.g., Vcnt1) is coupledin a feedback arrangement to the negative terminal of differentialamplifier 102. The output of replica charge pump 104 is fed back to thepositive terminal of the amplifier, and is also tied to the Vbn biasinputs of the replica charge pump and charge pump. The output of thedifferential amplifier is coupled to the Vbp bias inputs of charge pump106 and charge pump replica 104.

Charge pump 106 is responsive to signals applied to its UP and DOWNcharge pump inputs. As previously discussed, UP and/or DOWN signals maybe asserted in response to phase offset detection performed by phasedetector 30. Typically, assertion of the UP input causes a net positivecharge to be pumped to charge pump output (Vcnt1), thereby resulting ina higher control voltage or bias applied to delay lines 22 and 24, thuscausing a change in the relative delay between the lines (e.g., areduction in the delay). Delay line 22 typically has more activateddelay elements than line 24, so speeding up the individual delaycells/elements in each line causes a relative reduction in delay betweenthe two lines. Conversely, assertion of the DOWN input pumps a netnegative charge, producing a lower control voltage causing the bias tochange so that the delay of the lines is increased. This feedbackarrangement produces control in which the compared edges coming out ofthe delay lines tend toward and are maintained in a state of alignment.

As previously discussed, phase detector 30 is typically configured tohave no dead band, such that when the compared edges received at thephase detector inputs are aligned (or nearly so, such as within apre-defined range or percentage of the reference period), both the UPand DOWN outputs are asserted briefly each cycle. In such a case, chargepump system 40 is configured so that substantially no net charge ispumped from the charge pump output. To achieve this, the charge pumpmust be appropriately biased at bias inputs Vbp and Vbn to ensure thatsubstantially no net charge is pumped upon simultaneous assertion of theUP and DOWN signals at the inputs of charge pump 106. The feedbackcoupling of charge pump 106 and charge pump replica 104, along withoperation of differential amplifier 102, causes the charge pump systemto automatically and dynamically self-bias. As will be shown, thisdynamic self-biasing operation produces a bias condition (e.g., asetting of bias levels Vbp and Vbn) of the charge pump which causes thecharge pump to produce no net charge during simultaneous assertion ofthe UP and DOWN signals.

Referring now to FIG. 5, an exemplary embodiment of charge pump 106 isdepicted. The exemplary charge pump includes PMOS transistors 110, 112,113, 114, 116 and NMOS transistors 120, 121, 122, 124 and 126.Transistors 110 and 112 are configured to operate in saturation mode ascurrent sources biased by Vbp. Transistors 124 and 126 are alsoconfigured to operate in saturation mode as current sources, but biasedby Vbn. Between biased current sources 112 and 126 are two parallelbranches, the first being a series transistor stack includingtransistors 114 and 121, with the other branch being a series stackincluding transistors 116 and 122. Between biased current sources 110and 124 is a series stack including transistors 113 and 120. Transistors113, 120, 114, 121, 116 and 122 are operated as switches, with thefollowing gate connections:

transistor 113: Vss

transistor 120: Vdd

transistor 114: UP (e.g., from phase detector)

transistor 121: DOWN inverted (e.g., from phase detector)

transistor 116: UP inverted

transistor 122: DOWN

The output of the charge pump is taken between devices 116 and 122.Charge is pumped to/from the output based on bias levels Vbp and Vbn,and in response to the state of the UP and DOWN signals. The accumulatedcharge at the charge pump output (e.g., as stored at loop filtercapacitor 42) may be referred to as the control voltage Vcnt1.

For example, upon assertion of UP (with DOWN remaining unasserted),transistor 116 is switched on to conduct, while transistor 122 remainsoff. A positive net charge driven by current source 112 is then pumpedto the output. As previously discussed, this increased charge canproduce an increased control voltage to vary (e.g., decrease) therelative amount of signal delay in delay lines 22 and 24.

Similarly, upon assertion of DOWN (with UP remaining unasserted),transistor 116 remains off, and transistor 122 is switched on toconduct. Biased current source 126 then draws charge off from the output(i.e., a negative net charge is pumped). The resulting decreased chargeat the output may be used to slow down the delay lines and therebyincrease the relative amount of delay between the lines (e.g., to bringthe signals into alignment).

Upon simultaneous assertion of both UP and DOWN, current is permitted toflow through transistors 116 and 122. As previously discussed, thecharge pump system may be configured to dynamically generate biasinglevels Vbp and/or Vbn so as to equalize currents flowing throughtransistors 116 and 122. In this bias condition, substantially no netcharge is pumped to the charge pump output, thereby reducing oreliminating static phase offset in the alignment of signals coming outof delay lines 22 and 24.

The same bias condition that causes simultaneous assertion of UP andDOWN to pump no net charge through OUT also causes net 125 to settle ata voltage approximately equal to Vcnt1. Transistors 113 and 120 are bothalways on, and transistors 110 and 124 are biased to produce the samecurrents. Due to the finite impedance of current sources 110 and 124,net 125 settles to a voltage approximating Vcnt1.

When UP is deasserted, current from transistor 112 is switched to net125, where it is harmlessly dumped through transistors 120 and 124.Similarly, when DOWN is deasserted, current from transistor 126 is alsodumped into net 125. Because one of transistors 114 and 116 is alwayson, and one of transistors 121 and 122 is always on, transistors 112 and126 have constant current flow to relatively unvarying voltages, so thatthe response of the charge pump to the pulse width of the UP and DOWNsignals is more nearly linear. Better linearity reduces jitter while inlock.

Referring now to FIG. 6, an exemplary embodiment of replica charge pump104 is depicted. Replica charge pump 104 includes a PMOS current sourcetransistor 130 biased by Vbp; a PMOS switch transistor 132 with its gatetied to Vss; an NMOS switch transistor 134 with its gate tied to Vdd;and an NMOS current source transistor 136 biased at its gate by Vbn.

Typically, transistor devices 130, 132, 134 and 136 are matched totransistor devices 112, 116, 122 and 126, respectively, of charge pump106. For example, the transistor devices may be of the same type,width/geometry, etc. Charge pump replica 104 thus replicates a currentpath of charge pump 106, namely, the path through transistors 112, 116,122 and 126 when the UP and DOWN signals are simultaneously asserted.

Amplifier 102 may be implemented in a variety of different ways.Typically, amplifier 102 itself is self-biased, with its operatingcurrent being derived from its output voltage in order to cancel out itsinput offset. FIG. 7 depicts one exemplary embodiment. As indicated, theamplifier output is coupled to the Vbp bias node (FIGS. 4-6). Thenegative amplifier input is feedback-coupled with the charge pump outputVcnt1, while the other input (positive) is feedback-coupled with theoutput of replica charge pump 104 and NMOS bias voltage Vbn. This is butone exemplary embodiment. According to various alternate embodiments,additional differential amplifiers are employed (e.g., to feed NMOS biasinput Vbn, by comparing the charge pump output with the PMOS biasvoltage Vbp), in addition to or instead of amplifier 102.

The feedback topology of charge pump 106, charge pump replica 104 anddifferential amplifier 102 automatically and dynamically biases thecharge pump current sources, typically without reference to any externalfixed level. For example, if Vbn is less than Vcnt1, the output voltageof differential amplifier 102 will drop, thus reducing the PMOS biasvoltage Vbp further below Vdd until the output of the charge pumpreplica rises to equal Vcnt1. Conversely, if Vbn is greater than Vcnt1,the output of differential amplifier 102 will rise until the feedbackmechanism produces a PMOS bias voltage Vbp at which the charge pumpreplica output is equal to the stored charge pump output Vcnt1. In otherwords, the amplifier adjusts the PMOS bias voltage Vbp so that thediode-connected NMOS device (e.g. transistors 134 and 136, because Vbnis tied to OUT) produces Vcnt1 at its gate and drain.

The charge pump replica is thus automatically and dynamically biased sothat its output tracks the Vcnt1 output of charge pump 106. Also, aspreviously discussed, the charge pump replica is configured, via devicematching and the tied gate voltages of transistors 132 and 134, toreplicate the current pathway of charge pump 106 when both the UP andDOWN signals are asserted. Since the corresponding devices (currentsources) of charge pump 106 are commonly biased, the charge pump isautomatically and dynamically biased to have its output remainsubstantially constant (i.e., not net charge pumped) when both the UPand DOWN signals are asserted. In other words, when the UP and DOWNinputs of the actual charge pump are both asserted, it will mirror theoperation of charge pump replica 104 and produce no net charge at itsVcnt1 output. Accordingly, control is exerted over edge alignmentsduring lock, and/or within a pre-defined dead band range, in order tominimize or eliminate static phase offset and wandering alignment whenthe DLL system is in lock.

As previously indicated, the current source transistors (110, 112, 124,126, 130 and 136) typically are operated in the saturation region. Eachcan thus function as a current source independent of common-mode voltageexcursions on the nodes above/below.

Within amplifier 102, transistors 160 and 162 (FIG. 7) typically will beoperated in saturation as well, to provide maximum gain. Increased gainincreases response time of the feedback system, so that the replicafeedback biased charge pump minimizes the static phase offset of theDLL.

Referring still to exemplary amplifier 102 (FIG. 7), NMOS bias voltageVbn is tied to transistor 160, which is in series with transistor 164.Let us suppose that the gate-to-source voltage of amplifier transistors160, 162 and 164 is equal to Vgs when the inputs of the amplifier areequal, since transistor 164 is twice the size of transistors 160 and162. In order to prevent device 164 from entering linear region, thevoltage at the node above the transistor should be at least Vgs−Vt. Itcan be set to exactly Vgs−Vt with the correctly established currentdensities. This relationship may be converted into a constraint on therelative transistor sizing, as follows:

Vbn=Vgs+(Vgs−Vt)

Vbn−Vt=2*(Vgs−Vt)

(Vbn−Vt)̂2=4*(Vgs−Vt)̂2

When a transistor is in saturation, current density (current/devicearea) is proportional to gate-to-source voltage, minus the thresholdvoltage, squared:

J=˜(Vgs−Vt)̂2

Thus, in the present example, the current density in transistors 164,160 and 162 must be ¼ of the current density in transistors 124, 126 and136, which are all biased by Vbn.

This current density ratio may be assured by the relative sizes ofdevices 112, 166, 168, 160, 162 and 164. First, note that the currentthrough device 112 will equal the current through device 126 when thecharge pump is not pumping charge. Because devices 112 and 166 both actas PMOS current sources in saturation region, are both biased by Vbp,and have a 4:1 width ratio, the current through 166 will be ¼ of thecurrent through 112. Devices 168 and 164 act as an NMOS current mirror,so that the current through device 164 will be twice that through device168, and thus half of that through device 126. The current throughdevice 164 splits in two, so that the current through devices 160 and162 is one-fourth of that through device 126.

Thus devices 164, 160 and 162 have one-fourth the current density ofdevice 126, so that all devices will stay in saturation in all biasconditions, which is desired in the present example.

The above-described amplifier configuration is but one example of aspecific implementation. Many other configurations may be employed inthe feedback self-biasing arrangement described above.

FIG. 8 shows an exemplary delay element 200 in a delay line 202according to the present description. The delay lines of the presentdescription typically include multiple delay elements 200. In thepresent example, the delay element includes a plurality of logic gates(NAND gates) which serve to delay signals, such as a reference clock,passing through the delay line. The logic gates are powered by uppervoltage VVdd and lower or substrate voltage Vss. VVdd may be a regulatedsupply voltage, as explained in more detail below.

The delay elements 200 are enabled via a select input, such as from adecoder, as described above with reference to FIG. 3. In the presentembodiment, the delay element after the one with the wrap leg selectedalso has the wrap leg selected, so as to force a high into the outputchain input of the first selected wrap delay element. The status of theselect inputs determines the length of the input chain and output chain(i.e., the number of delay elements the signal passes through). In afirst state, the select input will enable the delay element, so that thepropagating signal continues through gate 210 to the right along theinput chain (to the next delay element). In the other state, the signalis wrapped down through gates 212 and 214 to the output chain, and thenpropagates to the left through all of the delay elements to the leftthat the signal passed through during its propagation along the inputchain. Thus, the signal in the delay line may pass along an input chaincontaining N delay elements, and then at the N+1 delay element, theselect signal may cause the signal to wrap around and then propagateback through gate 214 of each prior delay element.

The select signals from the decoder thus determine the number ofactivated delay elements through which the signal travels. As discussedin more detail below, the speed of each delay element is determined inpart by the regulated supply voltage VVdd. The delay of the delay lineis thus a function of the regulated supply voltage and the number ofactivated elements in the line.

Dummy loads may be employed within delay line 202. In the presentexample, dummy loads are implemented within each delay element as gates220 and 222. This may be employed to ensure that every NAND gate in thedelay line sees a substantially identical load. Thus, no matter how manydelay stages/elements are selected, both rising and falling input edgestravel through an equal number of rising and falling transitions throughidentically-loaded NAND gates. In certain applications, this propertyreduces pulse width distortion and improves linearity of delay withrespect to number of stages selected.

Referring now to FIG. 9, certain CMOS NAND gates have asymmetricalinputs. Because the signal passes through the wrap (upper) leg of NANDgate 401 in just one location on the delay line (i.e. the delay elementat which the signal is wrapped to the output chain), if a standard CMOSNAND were used here it would cause at least one gate in the path to seea different loading and delay environment than the rest, which couldintroduce pulse width distortion. Instead, flip-stack NANDs may be used,as indicated in FIG. 9. The symmetrical inputs enable every gate in thechain to see substantially identical loading and delay. Use of asymmetrical logic gate such as a flip-stack NAND may also minimize thedelay in each element (thus providing higher resolution), and may avoiddistorting rising and falling edges differently.

In order to maximize resolution, it may be desirable to minimize thedelay of each delay line element 200. It is also desirable in many casesto maintain equal delays for rising and falling edges. To ensureconstant current dissipation, the delay line stage selected to wrap theedge to the output chain must also prevent the edge from traveling anyfurther along the input chain. Otherwise there would be two edges in thedelay line simultaneously and current dissipation would double.According to one example, a CMOS NAND is employed to rapidly MUX thewrapped edge and kill the forward edge. By using the same gate for bothfunctions, the delay line can avoid propagating rising and falling edgesthrough different circuit structures, and thus distorting the two edgesdifferently. The resulting delay stage is inverting along both the inputchain and output chain. Because each additional selected stage adds twoinverting gates to the chain, the overall delay line is alwaysnoninverting.

Referring now to FIG. 10, further exemplary aspects of delay line 202will be described. It will be appreciated that the delay line featuresdiscussed herein may be employed in connection with the above-describedmaster or slave delay lines, or with other delay lines in a DLL system.Delay line 202 may include a plurality of delay elements, which may besimilar to those described above with reference to FIG. 8. The delayelements are driven by supply voltages Vss and regulated supply voltageVVdd. The input signal is applied to the beginning of the chain of delayelements. The signal passes through a number of elements which dependson the status of the select signals (not shown in FIG. 10), and thenpasses through the output of the chain of delay elements. Additional,selective delay may be added via additional delay elements 240. Theinitial output from the first chain of delay elements is designated inthe figure as “outc,” with the ultimate output of the delay line beingdesignated as “output.”

PMOS device 242 acts as a delay regulator to control a current/voltageapplied to delay elements 200. More particularly, device 242, acting asa current source, controls VVdd supply current directly, and because thedelay line acts as a constant impedance, device 242 controls the voltageon VVdd as well. This, in turn, controls the speed of the logic gatesand thus the amount of signal delay for the delay element. As shown inFIG. 10, a single biased current source may be used to supply operatingcurrent/voltage to all of the delay elements 200. The bias input todevice 242 typically is generated by bias generator 44, in response tooutput from charge pump system 40 and filter 42.

Device 242 acts as a biased current source to node VVdd. If the circuitsupplied by VVdd dissipates current intermittently (for example, onlywhile an edge is propagating through the delay line), current source 242must turn on and off. The transitions between the on and off states arenot instant and will affect the delays of the first and last delayelements in ways that may be sensitive to supply and substrate noise.

In order to reduce this sensitivity, it is desirable to cause the delayline elements supplied by VVdd to dissipate constant current, andpresent a constant impedance. However, CMOS gates typically dissipatecurrent only when their output changes state. Thus a line of delayelements using CMOS gates dissipates current only when an edge is inflight within the line of elements.

Accordingly, a current dissipation circuit, such as current dump 244,may be implemented. Current dump 244 is configured to draw current fromdelay regulator 242 when an edge is not in flight within the line ofdelay elements 200. FIG. 11 depicts an exemplary implementation of acircuit for detecting when an edge is in flight within delay elements200, and for drawing current from device 242 if no edge is in flight.

Referring particularly to FIG. 11, an XOR implementation may beemployed, in which current dump 244 only draws current if no edge ispresent within the line of delay elements 200. In particular, if theinput and OUTC of delay elements 200 have different values (e.g., one ishigh and the other low), then it is determined that an edge is in flightor present within the line of delay elements 200. Accordingly, logicgates within the line of delay elements are drawing current from device242. In this state, neither of the transistor stacks shown in FIG. 11are conducting, and current is thus not drawn by current dump 244 fromdelay regulator 242.

Conversely, if the input and OUTC are at the same level (no edge inflight), the XOR implementation of FIG. 11 enables a circuit mirroringthe logic gates that sinks a constant current through one of thetransistor stacks which replicates the current sunk by the delay linewhen an edge is in flight. With this current compensation circuit, thecurrent dissipation is more nearly constant, thus the voltage VVdd ismore nearly constant. This creates a more linear delay response andkeeps the semiconductor devices within delay elements 200 closer totheir optimal bias point. This also reduces supply and substrate noisesensitivity, and may eliminate or reduce jitter. Furthermore, a smallcapacitor 246 on VVdd can filter the very high frequency impedancevariation from successive gates switching, so that the impedance andthus voltage on VVdd remains more nearly constant. Thus, the delay ofthe various delay elements 200 within the delay line is better matched,and the delay of the delay line is less sensitive to supply andsubstrate noise, which will reduce jitter.

In addition, rather than the two current dump paths of FIG. 11, analternate implementation of the current dump could employ an explicitXOR gate enabling a single current dump path.

It will be appreciated that the resolution of a delay line is oftenlimited by the delay of each delay element. Finer resolution may beattained in any of the examples herein through interpolative methods.Specifically, finer resolution may be attained by using the outputs oftwo consecutive delay elements (e.g., two adjacent delay elements 200)and interpolating an output signal from these outputs.

In certain settings, it can be difficult to make a delay line thatdelays rising and falling edges identically. If the edge types aredelayed differently, the delay lines may distort the pulse width of theinput signal, which can be undesirable. To correct this problem, asecondary control loop may be added, as shown in FIG. 12.

The DLL of FIG. 12 includes a primary or main control loop as previouslydescribed, including phase detector 30, charge pump system 40, filter 42and bias generator 44. The secondary control loop includes phasedetector 30 a, charge pump system 40 a, filter 42 a and bias generator44 a. The components of the secondary control loop operate similarly tothose of the main control loop but on the opposite edge type, asindicated by the inverters on the inputs of phase detector 30 a. Forexample, the primary loop may be configured to perform phase detectionand delay control on rising edges from the delay lines, with thesecondary control loop operating on falling edges.

Except as otherwise indicated or described, the components of thesecondary control loop operate similarly to those of the master controlloop.

The secondary control loop includes a phase detector 30 a that receivesan inverted reference clock signal (via delay line 24) and the output ofdelay line 22. Typically, a inverted reference edge is compared againstthe previous inverted reference edge after being delayed by a fullreference cycle through delay line 22.

Charge pump system 40 a receives up and down outputs from phase detector30 a and produces a charge output proportional to the pulse width on theup or down outputs for loop filter 42 a. Other schemes might produce afixed charge rather than a proportional charge to the phase differencebetween the reference and delay line output signals.

A loop filter capacitor of loop filter 42 a may store the current extradelay value as a second control voltage. This voltage is proportional tothe time integral of the error charge signal from charge pump system 40a.

Bias generator 44 a converts the second control voltage into a secondbias level for the extra adjustment to be applied to edges of a specifictype (falling edges in the present example). Secondary adjustment may beapplied to one or more delay elements in the input or output of thedelay chain, as in the preferred embodiment. They can also be applied toall alternating gates (not delay elements) of the delay chain, bycoupling every other gate to a secondary current source VVdd2. In thislatter case, the secondary control loop can operate on delayed fallingedges through the two delay lines. In such a case, the falling edgeadjustment circuit will have more of an impact on the 360° delay linedue to the larger number of selected secondary adjustment stages.

However, in the former case, both delay lines will adjust falling edgesthe same way which will likely lead to control problems in the secondaryloop as the loop gain will be small and possibly negative. This issuemay be resolved, as shown in FIG. 13 (depicting an alternateimplementation of a portion of FIG. 12), by inverting the edge at theinput of the 0° delay line and inverting again after the 0° delay line,and by ensuring that the bandwidths of the primary and secondary controlloops are significantly different. According to one example embodiment,the primary loop has higher bandwidth, in which case the primary controlloop will still operate as described above since it sees the fallingedge delay adjustment as relatively constant, and will adjust to a lockpoint given the state of the falling edge correction circuit.

Note that the inputs to the secondary phase detector in FIG. 13 areopposite those of the primary phase detector, so that the secondaryphase detector triggers off edges derived from falling refclk edges.

With this configuration, the falling edge adjustment will only beobserved by the secondary control loop in the 0 deg delay line so thatthe gain of the secondary control loop will be well defined. It shouldbe appreciated that the added inverters in the bath of both delay linesshould be matched so that their added delays are identical.

This second bias level is indicated as “secondary bias” in FIG. 10. Asindicated, the secondary bias level may be applied to alternate ones ofdelay elements 240, to adjust the delay of the falling input edges morethan the delay of rising input edges, in response to the secondary biaslevel generated by the secondary control loop.

It will be appreciated that the embodiments and method implementationsdisclosed herein are exemplary in nature, and that these specificexamples are not to be considered in a limiting sense, because numerousvariations are possible. The subject matter of the present disclosureincludes all novel and nonobvious combinations and subcombinations ofthe various features, functions, and/or properties disclosed herein. Thefollowing claims particularly point out certain combinations andsubcombinations regarded as novel and nonobvious. These claims may referto “an” element or “a first” element or the equivalent thereof. Suchclaims should be understood to include incorporation of one or more suchelements, neither requiring nor excluding two or more such elements.Other combinations and subcombinations of the disclosed features,functions, elements, and/or properties may be claimed through amendmentof the present claims or through presentation of new claims in this or arelated application. Such claims, whether broader, narrower, equal, ordifferent in scope to the original claims, also are regarded as includedwithin the subject matter of the present disclosure.

What is claimed is:
 1. A delay-locked loop, comprising: a phase detectorconfigured to receive two signals, one of the signals being delayedrelative to the other of the signals, the phase detector having an UPoutput and a DOWN output; and a charge pump system operatively coupledwith the phase detector and including: a charge pump configured to beresponsive to assertion of actuating signals from the UP output and theDOWN output so as to control pumping of charge from the charge pumpsystem, such pumped charge being usable to control a delay line carryingone of the two signals, so as to control relative delay occurringbetween the two signals; and a feedback control loop configured todynamically adjust at least one bias signal at the charge pump so as tominimize net charge pumped from the charge pump system duringsimultaneous assertion of actuating signals from the UP output and theDOWN output.
 2. The delay-locked loop of claim 1, where the phasedetector is coupled to two delay lines, with the delay-locked loop beingconfigured to lock the relative delay between the two delay lines to onereference cycle.
 3. The delay-locked loop of claim 1, where the chargepump includes: a first switch coupled with a first current source andconfigured to be actuated via assertion of an actuating signal from theUP output; and a second switch coupled with a second current source andconfigured to be actuated via assertion of an actuating signal from theDOWN output, the charge pump being adapted so that biasing of thecurrent sources and actuation of the switches controls pumping of chargefrom the charge pump system.
 4. The delay-locked loop of claim 3, wherethe feedback control loop includes a charge pump replica, the chargepump replica having: a first switch coupled with a first current source,the first switch and the first current source of the charge pump replicabeing matched to the first switch and the first current source of thecharge pump; and a second switch coupled with a second current source,the second switch and the second current source of the charge pumpreplica being matched to the second switch and the second current sourceof the charge pump.
 5. The delay-locked loop of claim 4, where the firstcurrent source of the charge pump and the first current source of thecharge pump replica are commonly biased, and where the second currentsource of the charge pump and the second current source of the chargepump replica are commonly biased.
 6. The delay-locked loop of claim 4,where the first switch and the second switch of the charge pump replicaare tied closed.
 7. The delay-locked loop of claim 6, where an output ofthe charge pump is fed back through a differential amplifier to bias atleast one of the first current sources and the second current sources,thereby dynamically generating a bias condition in which an output ofthe charge pump replica converges to an output of the charge pump. 8.The delay-locked loop of claim 1, further comprising first and secondcontrol loops, the first control loop being configured to controlrelative delay occurring between the two signals based on phase offsetdetection between rising edges of the two signals, the second controlloop being configured to control relative delay occurring between thetwo signals based on phase offset detection between falling edges of thetwo signals.
 9. A delay-locked loop, comprising: a first delay line; asecond delay line; a phase detector coupled to outputs of the first andsecond delay lines, the phase detector including an UP output and a DOWNoutput; a charge pump coupled to the phase detector so that the UPoutput of the phase detector is coupled to a first charge pump input andso that the DOWN output of the phase detector is coupled to a secondcharge pump input, where the charge pump is configured to pump chargevia a charge pump output based on a bias condition and on levels at thefirst charge pump input and the second charge pump input; and a chargepump replica coupled with the charge pump and configured to dynamicallyadjust the bias condition so that, in response to an assertion conditionat the first charge pump input and the second charge pump input, thecharge pump pumps a net charge that tends toward a desired valueassociated with such assertion condition.
 10. The delay-locked loop ofclaim 9, where the assertion condition is a simultaneous assertion ofactivating signals at the first charge pump input and the second chargepump input.
 11. The delay-locked loop of claim 10, where the desiredvalue is substantially zero net charge.
 12. The delay-locked loop ofclaim 9, where the charge pump and the charge pump replica each have afirst common bias input and a second common bias input.
 13. Thedelay-locked loop of claim 12, where the charge pump output and a chargepump replica output are coupled to a differential amplifier, and wherean output of the differential amplifier biases at least one of the firstcommon bias input and the second common bias input.
 14. The delay-lockedloop of claim 12, where the charge pump output and a charge pump replicaoutput are operatively coupled to the first common bias input and thesecond common bias input.
 15. The delay-locked loop of claim 9, furthercomprising first and second control loops, the first control loop beingconfigured to control relative delay occurring between the first andsecond delay lines based on phase offset detection between rising edgesoutput from the first and second delay lines, the second control loopbeing configured to control relative delay occurring between the firstand second delay lines based on phase offset detection between fallingedges output from the first and second delay lines.